1. Field of Invention
The present invention relates generally to an integrated circuit. More particularly, the present invention relates to a structure of a metal-oxide-semiconductor field effect transistor (MOSFET) and a fabrication method thereof.
2. Description of Related Art
For high-frequency applications of the MOSFET, e.g. radio frequency (RF) communication and high-speed analog and digital integrated circuits, effects of the internal parasitic capacitances of the MOSFET must be taken into account. Internal parasitic capacitances of the MOSFET include the gate-to-drain overlap capacitance, which arises because the drain region extends slightly under the gate electrode.
The effects of the internal capacitances are analyzed herein with reference to FIGS. 1A-1D. FIG. 1A illustrates the MOSFET common-source amplifier configuration. Three large-valued capacitors C1, C2, and C3 are used to couple the gate G, source S and drain D of a MOSFET Q to signal source Vi in s-domain, ground 100 and load resistance RL, respectively. This amplifier has an output voltage Vo in s-domain. The signal generator generating the signal source Vi has a resistance Rs. A dc current source I is used to bias the MOSFET Q, and is connected to a negative supply voltage −Vss. A large resistor RG connects the gate G to ground 100, and a resistor RD connects the drain D to a positive supply voltage VDD. It's assumed that the source S of the MOSFET Q is connected to the substrate, and in the following analysis of the high-frequency response of the MOSFET common-source amplifier of FIG. 1A these capacitors C1, C2, and C3 act as perfect short circuits.
FIG. 1B illustrates a small-signal equivalent circuit of the MOSFET common-source amplifier of FIG. 1A. In FIG. 1B, a small-signal equivalent circuit model for the MOSFET is used to replace the MOSFET Q of FIG. 1A. The small-signal equivalent circuit model includes the gate-to-source parasitic capacitance Cgs with a voltage Vgs across its two terminals, gate-to-drain parasitic capacitance Cgd, a dependent current source gmVgs, and an output resistance ro, in which the MOSFET Q has a transconductance gm. The parasitic capacitance between the drain D and source S (or substrate) is usually neglected in an approximate analysis.
In most situations of interest, the MOSFET operates in the saturation region. When the MOSFET Q is in the saturation region, the parasitic capacitance Cgs includes the gate-to-channel capacitance, the gate-to-source overlap capacitance and the gate-to-substrate parasitic capacitance. The gate-to-channel capacitance is the major component of the parasitic capacitance Cgs. The parasitic capacitance Cgd is entirely an overlap capacitance between the drain D and the gate G, and has a typical value of 1 to 10 fF (femto-Farad). As shown in FIG. 1B, the output resistance ro, resistor RD and load resistance RL are combined to be an equivalent resistance R′L.
FIG. 1C illustrates a simplified version of the small-signal equivalent circuit of FIG. 1B. A Thevenin voltage ViRG/(RS+RG) and a Thevenin resistance R′ (equal to RS in parallel with RG) are obtained by applying Thevenin's theorem at the input side of the circuit of FIG. 1B. Since the overlap capacitance Cgd is small, the current through it is very small and thus can be neglected in determining the output voltage Vo. Therefore, the output voltage Vo can be expressed asVo≈−gmVgsR′L
FIG. 1D illustrates the input (gate) side circuit of FIG. 1C after replacing the overlap capacitance Cgd with the equivalent Miller capacitance CM at the input side between the gate G and ground 100. Using the ratio of the voltages at the two sides of the overlap capacitance Cgd of FIG. 1CVo/Vgs=−gmR′Lenables us to find the equivalent Miller capacitanceCM=Cgd(1+gmR′L)
With reference to FIG. 1D, the parasitic capacitance Cgs and the equivalent Miller capacitance CM are in parallel, so they can be combined to be an equivalent capacitance CT. The input side circuit in FIG. 1D, an input RC circuit, is a circuit of a first-order low-pass filter whose time constant is CTR′. This first-order circuit determines the high-frequency response of the common-source amplifier of FIG. 1A, introducing a dominant high-frequency pole. The dominant high-frequency pole represents the upper 3-dB frequency ωH which isωH=1/CTR′Thus the high-frequency gain AH of the common-source amplifier can be expressed asAH=AM(1/[1+s/ωH])where s is the complex frequency, and AH is the midband gain. The high-frequency response analysis described above can be found and is explained in more detail in “Microelectronic Circuits”, International Thomson Publishing, 3 ed, chapter 7, by Adel S. Sedra and Kenneth C. Smith.
According to the above analysis, the overlap capacitance Cgd plays an important role in determining the high-frequency response. The overlap capacitance Cgd affects the equivalent capacitance CT, thereby affecting the upper 3-dB frequency ωH and the high-frequency voltage gain AH. This is the Miller effect. If the overlap capacitance Cgd is reduced, the upper 3-dB frequency ωH and the high-frequency gain AH can be increased. On the other hand, since the parasitic capacitance Cgs is an important factor affecting the MOSFET device parameters, including the threshold voltage Vt and the drain-to-source current IDS, and thus affecting device performance, the gate-to-source overlap capacitance shouldn't be reduced.
Accordingly, there is a need for a MOSFET fabricating method that can be used to reduce the gate-to-drain overlap capacitance Cgd, and thus to improve the high-frequency response of MOSFET amplifiers.